1. Field and Background of the Invention
High speed downlink packet access (HSDPA) described in 3GPP TR25.858 V5.0.0 (2002-3), “High speed downlink packet access: Physical layer aspects (Rel5)”, also known as Rel5, is underway supporting the evolution of third-generation systems to meet the rapidly developing needs on high data rate. Various technologies are considered in HSDPA proposals such as adaptive modulation and coding (AMC), hybrid automatic repeat request (HARQ), fast cell selection (FCS), multiple input multiple output (MIMO) antenna processing and multicode transmission. HSDPA user equipment (UE) is suffering from multiple access interference (MAI) induced by its own multiple spreading codes and speech user interference (SUI) induced by a co-existing interfering speech user signal also known, for example, as Rel99 according to 3GPP TR25.101, “UE Radio Transmission and Reception (FDD)”.
A comprehensive multiuser detection methods to suppress interference can be found in S. Verdú, Multiuser Detection: Cambridge University Press, 1998. Most of them are proposed for uplink communications with the knowledge of all the spreading codes. The HSDPA UE only knows its own spreading codes in multicode transmission, allocated power and modulation alphabet and has no knowledge of the interfering speech user signal. That is why blind speech user interference cancellation for suppressing the SUI is a major challenge.
Blind multiuser detectors require no training data sequence, only the knowledge of the desired user spreading code. A blind adaptive MMSE (minimum mean-square error) multiuser detector is introduced by M. Honig, U. Madhow, and S. Verdú “Blind adaptive multiuser detection,” IEEE Trans. Inform. Theory, vol. 41, pp. 944-960, July 1995. A subspace approach for blind multiuser detection is presented by X. Wang and V. Poor, “Blind multiuser detection: a subspace approach,” IEEE Trans. Inform. Theory, vol. 44, pp. 677-690, March 1998, where both the decorrelating and the MMSE detector are obtained blindly. A blind solution based on higher order statistics and nonlinear cancellation is presented by D. Samardzija, N. Mandayam, and I. Seskar, “Nonlinear adaptive blind interference cancellation for DS-CDMA systems,” in The IEEE Vehicular Technology Conf.e (VTC), Boston, Mass., September 2000. Alternative adaptive and blind solutions have been analyzed by S. Ulukus and R. Yates, “A blind adaptive decorrelating detector for CDMA systems,” IEEE J. Select. Areas Commun., vol. 16, pp. 1530-1541, October 1998, and overviewed by U. Madhow, “Blind adaptive interference suppression for direct-sequence CDMA,” in Proc. IEEE, Special Issue on Blind Identification and Equalization, October 1998, pp. 2049-2069.
These proposed adaptive receivers are based mostly on the linear MMSE criterion. Through the central limit theorem, the SUI tends to be a Gaussian random process when the number of interfering speech users with random time delay is high enough. In this case, the MMSE principle leads to minimization of error probability. However, most of the base station (BS) transmission power and spreading codes will be assigned to a high speed user in HSDPA with a reasonably lower number of speech users in practical synchronous DL transmission. In this situation, the SUI does not tend to be Gaussian distributed, which triggers the research and development on SUI detection and nonlinear interference cancellation.
2. System Model
The discrete-time received signal at HSDPA UE can be presented asr=H(ShAhbh+SsAsbs+spAp bp)+n ∈ CP×1   (1)wherein H ∈ CP×P is a matrix containing multipath channel impulse response in a time domain, P=Rchip×TTTI is a number of chips per transmission time interval (TTI) with Rchip=3.84 Mcps (typical value) is the chip rate and TTTI=2 ms (typical value) is the time period of TTI, C is a complex space, n ∈ CP×1 is a noise vector.
      S    h    =            diag      (                                                  S              ^                        h                    ,          …          ⁢                                          ,                                    S              ^                        h                                    ︷                      P            /                          SF              1                                          )        ∈                  ⁢          ℜ              P        ×                  L          h                    is a block-based diagonal matrix over one TTI, wherein SF1=16 (typical value) is a spreading factor (SF) of a desired HSDPA signal with known spreading codes,
      L    h    =            P              SF        1              ⁢          N      h      is a number of HSDPA parallel data symbols transmitted per TTI, Nh is a number of assigned multicodes,  is a real value space, Ŝh=[s1h;s2h; . . . ;sNhh]∈ SF1×Nh  is a spreading code matrix over one HSDPA symbol period, sih ∈ SF1×1 is an ith assigned SF1-bit Walsh code vector, Ah ∈ Lh×Lh is a diagonal matrix with HSDPA symbol energy, bh ∈ CLh×1 is a transmitted HSDPA symbol vector per TTI.
      S    s    =            diag      (                                                  S              ^                        s                    ,          …          ⁢                                          ,                                    S              ^                        s                                    ︷                      P            /                          SF              2                                          )        ∈                  ⁢          ℜ              P        ×                  L          s                    is a block-based diagonal matrix over one TTI, wherein SF2=128 (typical value) is a spreading factor of an interfering speech user signal with unknown spreading codes,
      L    s    =            P              SF        2              ⁢          N      s      is a number of interfering speech user parallel data symbols transmitted per TTI, Ns is a number of co-existing interfering speech users, Ŝs=[s1s;s2s; . . . ;sN,s]∈ SF2×N, is a spreading code matrix over one interfering speech user symbol period, sis ∈ K2×1 is the ith assigned SF2-bit Walsh code vector, As ∈ Ls×Ls is a diagonal matrix with interfering speech user symbol energy, bs ε CLs×1 is a transmitted interfering speech user symbol vector per TTI.
      s    p    =            [                                                  s              ^                        p                    ,          …          ⁢                                          ,                                    s              ^                        p                                    ︷                      P            /                          SF              3                                          ]        ∈                  ⁢          ℜ              P        ×        1            is a spreading vector for a common pilot channel (CPICH), wherein SF3=256 (typical value) is a spreading factor of a pilot signal,
      L    p    =      P          SF      3      is a number of parallel pilots per TTI, ŝp ∈ SF3×1 is one assigned K3-bit Walsh code vector, Ap ∈ Lp×Lp is a diagonal matrix with pilot energy, bp ∈ CLp×1 is a pilot vector per TTI.
3. Conventional RAKE Receiver
The conventional RAKE receiver (RAKER) neglects the MAI induced by its own spreading codes and SUI induced by co-existing interfering speech users and CPICH interference so Equation 1 can be rewritten for the RAKER as followsrraker=HShAhbh+(IMAI+ISUI+ICPICH)+n,   (2)so that a hard-decision (HD) data estimated by RAKER can be written as{circumflex over (b)}dec(z)=dec(AhHShHHHrraker),   (3)Wherein dec( ) is a decision device based on modulation alphabets, z is a SD output of RAKER, and ( )H denotes a complex conjugate transpose operation.
Common Pilot cancellation in UE has recently gained attention for CDMA cellular networks and it has been shown that the network capacity can be significantly improved as described in 3GPP TR 25.991: Feasibility study on the mitigation of the effect of the common pilot channel (CPICH) interference at the user equipment, 2002, and in 3GPP R4-01-1232, Motorola, “CPICH Cancellation Complexity.”
Since the UE has the knowledge of the CPICH on power, spreading codes and pilot symbols for channel estimation, the interference induced by the CPICH can be subtracted directly as follows{tilde over (r)}=r−ICPICH=r−HSpApbp.   (4)
Then the same RAKER principle of Equation 3 can be applied for the received signal with CPICH interference cancellation described by Equation 4.
4. Conventional PIC Receiver
In contrast to the RAKE receiver, a conventional parallel interference cancellation (PIC) receiver described by M. K. Varanasi and B. Aazhang, “Multistage detection for asynchronous code-division multiple-access communications,” IEEE Transactions on Communications, COM-38(4), April 1990, suppresses the MAI induced by its own spreading codes with the knowledge of allocated power, assigned spreading codes and modulation alphabet. However, it still neglects SUI induced by the co-existing interfering speech users and CPICH interference. Using the conventional RAKER output as the initial estimates as {circumflex over (b)}pic(0)={circumflex over (b)}raker, the hard-decision data estimates of PIC at mth stage can be described as{circumflex over (b)}pic(m)=dec(z−F{circumflex over (b)}pic(m−1))   (5)wherein F=G−diag(G) is the off-diagonal matrix, diag( ) denotes diagonal elements of the matrix, G=AhHShHHHHShAh is the cross-correlation matrix.